To increase the amount of information that can be transmitted in a given bandwidth of the radio frequency (RF) spectrum (i.e., to increase what is referred to in the art as “spectral efficiency”), modern wireless communications technologies such as Wideband Code Division Multiple Access (W-CDMA) use non-constant envelope modulation schemes in which both the amplitude (i.e., envelope) and angle (i.e., phase or frequency) of the RF signal to be transmitted are varied.
To accommodate non-constant envelope modulation schemes in conventional quadrature-modulator-based transmitters, the output power of the quadrature-modulator-based transmitter's RF power amplifier (PA) must be backed off to prevent the PA from clipping the signal peaks of the non-constant envelope signals. Maintaining linearity also requires that the PA be configured to operate exclusively in its linear region of operation (i.e., that a ‘linear PA’ be used). Failure to back off the output power and use a linear PA results in signal distortion at the output of the PA, making it difficult to comply with noise level restrictions imposed by communications standards.
The need to back off output power and use a linear PA results in a sacrifice of energy efficiency for linearity. This trade-off is highly undesirable, especially in applications in which the transmitter is used in battery-powered communications devices, such as in cellular handsets. Fortunately, an alternative type of transmitter, commonly known as a polar transmitter, is available, which avoids the efficiency versus linearity trade-off.
FIG. 1 is a simplified drawing of a typical polar transmitter 100. The polar transmitter 100 includes a baseband processor 102; an amplitude modulation (AM) path including an AM path digital-to-analog converter (DAC) 104 and an amplitude modulator 106; a phase modulation (PM) path including a PM path DAC 108 and a phase modulator 110; a PA 112; and an antenna 114.
The baseband processor 102 is responsible for performing the signal processing functions of the polar transmitter 100, including generating digital amplitude and phase modulation data streams ρ and θ from a digital message to be transmitted and formatting the digital amplitude and phase modulation data streams ρ and θ according to a predetermined modulation scheme. The AM and PM path DACs 104 and 108 convert the digital amplitude and phase modulation data streams ρ and θ into analog amplitude and phase modulation signals AM(t) and PM(t).
The amplitude modulator 106 modulates a DC power supply Vsupply according to the amplitude variations in the analog amplitude modulation signal AM(t), to provide an amplitude-modulated power supply signal Vs(t) which is coupled to the power supply port of the PA 112. Meanwhile, the phase modulator 110 modulates an RF carrier signal according to the phase variations in the analog phase modulation signal PM(t), to provide a phase-modulated RF carrier signal which is coupled to the RF input port of the PA 112.
Because the phase-modulated RF carrier signal at the RF input port of the PA 112 has a constant envelope, the PA 112 can be configured to operate as a highly efficient nonlinear PA without concern for distorting the amplitude information, which as explained above is processed in the separate AM path. Typically, the PA 112 is implemented as a Class D, E or F switch-mode PA operating between compressed and cut-off states (i.e., in a “compressed mode”), so that the output power of the PA 112 is directly controlled and modulated according to the amplitude variations in the amplitude-modulated power supply signal Vs(t). By modulating the power supply port of the PA 112 in this manner, the amplitude modulation represented in the original digital amplitude modulation data stream p is restored at the output of the PA 112, to provide the desired amplitude- and phase-modulated RF carrier signal.
Some communications systems require RF transmitters of wireless communications devices to be capable of controlling RF output power over a wide dynamic range. For example, the 3rd Generation Partnership Project (3GPP) TS 25.101 specification, which standardizes the transmission and reception requirements of user equipment (UE) in Universal Mobile Telecommunications System (UMTS) cellular networks, requires RF transmitters of UE to be capable of controlling RF output power over a range of −50 dBm to +24 dBm (where 1 dBm corresponds to a 1 milliwatt reference). Controlling RF output power over such a wide dynamic range presents a difficult challenge. In a polar transmitter, controlling RF output power is especially difficult at the lower end of the RF output power range. At low RF output powers, when the magnitude of the amplitude-modulated power supply signal Vs(t) is low, leakage (i.e., feed-through) of the phase-modulated RF carrier signal from the RF input of the PA 112, through an input-output parasitic capacitance, to the RF output of the PA 112 occurs. The leaked signal has the effect of degrading the signal-to-noise ratio (SNR) at the RF output of the PA 112. The degraded SNR makes it difficult to comply with noise level restrictions set by standards, e.g., such as those set forth in Section 6 of the 3GPP TS 25.101 specification.
To overcome this problem and extend the effective power control range to lower RF output powers, the PA 112 of the polar transmitter 100 can be configured to operate in what is referred to in the art as “product mode” (or “multiplicative mode”) during times the polar transmitter 100 is to transmit below some predetermined RF output power (e.g., less than 0 dBm), and in compressed mode at higher RF output powers. Such an approach is described in U.S. Pat. No. 7,010,276 to Sander et al. In product mode the PA 112 operates in its deep triode region, where the RF output power of the PA 112 is determined by the product of the magnitude of the amplitude-modulated power supply signal Vs(t) and the magnitude of the phase-modulated RF carrier signal, rather than just by the magnitude of the amplitude-modulated power supply signal Vs(t), as in compressed mode. RF output power is controlled in product mode by manipulating the amplitude of the RF input drive to the PA 112 (i.e., the amplitude of the phase-modulated RF carrier signal) while the magnitude of the amplitude-modulated power supply signal Vs(t) is held at a fixed value. This allows the RF output power to be controlled without having to reduce the magnitude of the amplitude-modulated power supply signal Vs(t) to a level that would result in SNR degradation at the RF output of the PA 112.
In addition to requiring a wide dynamic range in RF output power control, wireless communications standards also often require an RF transmitter to control its RF output power in certain well-defined steps. For example, the 3GPP TS 25.101 specification requires RF transmitters of UE to be capable of incrementing and decrementing RF output power in discrete power steps of 1, 2 and 3 dB, in response to transmit power control (TPC) commands received from UMTS base stations. The specification further requires that the RF output power, following each power step, comply with certain power-step-size-dependent power control ranges (i.e., tolerances), as shown in the table in FIG. 2. For example, a TPC command of +1 requires that the RF output power of an RF transmitter following a power step size of +1 dB to be within +/− 0.5 dB of the target RF output power.
Designing a polar transmitter that is capable of complying with a +/− 0.5 dB power control range following a 1 dB power step size presents a difficult challenge, particularly when the power step involves a mode switch between compressed and product modes. Ideally, and as illustrated in FIG. 3A, the actual RF output power of the polar transmitter, when plotted against a range of commanded RF output powers, is continuous over the full dynamic range of RF output powers the polar transmitter can be commanded to operate, including the mode transition region between product and compressed modes. However, in real applications the RF output power curves for the product and compressed modes behave differently depending on temperature. As shown in FIG. 3B, at some temperatures a time-varying power discontinuity (i.e., power gap) occurs between the compressed and product mode RF output power curves. For some temperatures the power discontinuity is wide enough that the +/− 0.5 dB power control range following a 1 dB step in RF output power specified in the UMTS standard cannot be complied with.
Previous attempts to account for the temperature-dependent power discontinuity between compressed and product modes rely on a power detector configured between the RF output of the PA 112 and the baseband processor 102. The power detector is configured to measure the RF output power during times when a power change command involves a mode switch from product mode to compressed mode or vice versa. The measured RF output power is fed back to the baseband processor 102, which compares the measured RF output power to an expected RF output power to determine the difference (i.e., error) between the two. Based on the error, the baseband processor then adjusts power control settings to the polar transmitter's PA 112 to compensate for misalignment of the product and compressed mode power curves.
Reliance on a power detector and feedback to account for the temperature-dependent power discontinuity between compressed and product modes presents a number of drawbacks. First, the power detector and other supporting circuitry increase the hardware complexity of the polar transmitter 100. Second, utilizing the power detector renders the polar transmitter 100 a closed-loop system, requiring a fast response time, particularly during times when a power step involves a mode switch and correction for misalignment of the product and compressed mode power curves. The fast response time results in higher system clock rates, which not only increases the complexity of the baseband processor 102 but also undesirably results in higher energy consumption. Third, accurate compensation for misalignment of the product and compressed mode power curves require that the power detector be capable of detecting RF output powers to within a fraction of a dB over a wide dynamic range of RF output powers. Unfortunately, a power detector having these capabilities is difficult to design, even with the aid of factory calibration.
Considering the drawbacks and limitations of previous attempts at controlling RF output power in polar transmitters, it would be desirable to have methods and apparatus for controlling RF output power in polar transmitters over a wide dynamic range with the power control accuracy demanded by modern wireless communication standards.